A key and essential component of every radio frequency (RF) communications system is the RF transmitter. As shown in FIG. 1, an RF transmitter 100 generally comprises a baseband modulator 102, a frequency upconverter 104, a power amplifier (PA) 106, and an antenna 108. The purpose of the baseband modulator 102 is to generate a baseband signal s(t) that contains a message to be transmitted and which is formatted in accordance with a predetermined modulation scheme. The purpose of the frequency upconverter 104 is to upconvert the baseband signal s(t) to RF, so that the message is capable of being propagated through space (i.e., transmitted over the air) to a remote receiver. The PA 106 is employed to increase the power of the RF signal before it is radiated by the antenna 108, thereby compensating for attenuation of the RF signal as it is transmitted over the air to the remote receiver.
In modern RF transmitters, the message to be transmitted is included in a binary-source data stream. The baseband modulator 102 groups the data bits in the binary-source data stream into a sequence of N-bit words, where N is some positive integer, and maps the pattern of bits in each N-bit word to one of M=2N possible symbols. The M symbols define the modulation scheme being employed, and affect how the amplitude and/or angle of the RF carrier signal is varied (i.e., modulated) to carry the message in the original binary-source data stream to the remote receiver. By mapping each N-bit word to one of M possible symbols, N=log2 M bits can be transmitted in each symbol.
The symbols generated by the baseband modulator 102 comprise a sequence of weighted impulses. To limit the bandwidths of the impulses, the baseband modulator 102 is further configured to shape each impulse by a band-limiting pulse p(t).
Mathematically, the baseband signal s(t) can be expressed as:
            s      ⁡              (        t        )              =                  ∑        n            ⁢                          ⁢                        a          n                ⁢                  p          ⁡                      (                          t              -                              nT                s                                      )                                ,where n is a symbol index, an is the nth symbol in the sequence of symbols, p(t) is the pulse at time t, and Ts is the symbol period. Each an is either a real or complex number having one of M possible states. For example, in the quadrature phase-shift keying (QPSK) modulation scheme, M=4, and an is given by an=ejπ(2dn+1)/2, where dn is an integer selected from the set {0, 1, 2, 3}.
Because the baseband signal s(t) is in general a complex signal, it is usually expressed in terms of its in-phase (I) and quadrature (Q) components, i.e., as s(t)=I(t)+jQ(t), and the baseband modulator 102 is configured to generate separate pulse-shaped I and Q baseband signals for each of the I and Q channels. This is illustrated in FIG. 2, which is simplified drawing of a typical RF transmitter 200. The RF transmitter 200 comprises a baseband modulator 202; I-channel and Q-channel digital-to-analog converters (DACs) 204 and 206; a transmit local oscillator (Tx-LO) 208; a quadrature modulator 210; a PA 212; and an antenna 214. Due to its use of the quadrature modulator 210, the RF transmitter 200 is referred to in the description below as the “quadrature-modulator-based” transmitter 200.
The quadrature modulator 210 of the quadrature-modulator-based transmitter 200 includes an I-channel mixer 216, a Q-channel mixer 218, a ninety-degree phase shifter 220, and a combiner 222. The I-channel and Q-channel DACs 204 and 206 convert the pulse-shaped I and Q baseband signals from the baseband modulator 202 into analog I and Q baseband signals. The quadrature modulator 210 then upconverts the analog I and Q baseband signals to RF. Specifically, the I-channel mixer 216 mixes the analog I baseband signal with an RF carrier signal provided by the Tx-LO 208, while the Q-channel mixer 218 mixes the analog Q baseband signal with a ninety-degree phase-shifted version of the RF carrier signal produced at the output of the ninety-degree phase shifter 220, thereby producing upconverted I- and Q-channel RF carrier signals. The upconverted I- and Q-channel RF carrier signals are then combined by the combiner 222 to produce the desired modulated RF carrier signal, which is finally amplified by the PA 212 and radiated over the air to a remote receiver by the antenna 214.
The quadrature-modulator-based RF transmitter 200 is satisfactory for many applications. However, it is not the most desirable solution when used in communication systems employing non-constant envelope modulation schemes. Current and next generation mobile telecommunications systems commonly employ non-constant envelope modulation schemes to achieve higher data rates for a given bandwidth of the RF spectrum than can be realized using constant envelope modulation schemes. However, as explained below, their use in quadrature-modulator-based RF transmitters requires a sacrifice of energy efficiency.
A non-constant envelope modulation scheme produces a baseband signal that has a non-constant (i.e., time-varying) envelope. Consequently, and as illustrated in FIG. 3, the modulated RF carrier signal presented to the RF input RFin of the PA 212 also has a non-constant envelope. To prevent the PA 212 from clipping the signal peaks of the modulated RF carrier signal, the input signal power to the PA 212 must be reduced. This technique, known as power back-off, ensures that the PA 212 always operates in its linear region of operation, even during times when the magnitude of the modulated RF carrier signal is at its peak. Unfortunately, while power back-off helps to ensure linearity it also undesirably results in a reduction in energy efficiency.
The energy efficiency of an RF transmitter is determined in large part by how efficient the RF transmitter's PA is, since the PA is usually the dominant consumer of energy in the RF transmitter. The energy efficiency of the PA is determined by the ratio of the PA RF output power to the direct current (DC) power supplied to the PA from the RF transmitter's power supply. Consequently, when power back-off is employed the energy efficiency of the RF transmitter is reduced. The reduction in energy efficiency is most severe for signals that have a high peak-to-average ratio (PAR). Unfortunately, many modern non-constant envelope schemes produce signals having high PARs.
An RF transmitter having low energy efficiency is undesirable in most any circumstance. It is particularly undesirable when the RF transmitter comprises a battery-powered RF transmitter, such as used in a cellular handset, since the low energy efficiency results in shortened battery life. Fortunately, an alternative type of communications transmitter known as a polar transmitter is available which avoids the linearity versus energy efficiency tradeoff that plagues the quadrature-modulator-based transmitter 200. In a polar transmitter the amplitude information (i.e., the signal envelope) is temporarily removed from the non-constant envelope signal while the remaining signal, which has a constant envelope, is upconverted to RF. As explained in more detail below, the previously removed signal envelope is used to modulate the power supplied to the PA as the upconversion process takes place. Because the signal applied to the RF input of the PA has a constant envelope, a more energy efficient nonlinear PA can be used without the risk of signal peak clipping.
FIG. 4 is a drawing showing the salient elements of a typical polar transmitter 400. The polar transmitter 400 comprises a baseband modulator 402; a Coordinate Rotation Digital Computer (CORDIC) converter (i.e., rectangular-to-polar converter) 404; an amplitude path including an amplitude path DAC 406 and amplitude modulator 408; an angle path including an angle path DAC 410 and angle modulator 412; a PA 414; and an antenna 416. The purpose of the CORDIC converter 404 is to convert the digital rectangular-coordinate pulse-shaped I and Q baseband signals from the baseband modulator 402 to digital polar-coordinate amplitude and angle component signals ρ and θ. The amplitude and angle path DACs 406 and 410 convert the digital amplitude and angle component signals ρ and θ into analog amplitude and angle modulation signals. In the amplitude path, the amplitude modulator 408 then modulates a direct current power supply voltage Vsupply (e.g., as provided by a battery) by the amplitude information in the analog amplitude modulation signal. The resulting amplitude-modulated power supply signal Vs(t) is supplied to the power supply port of the PA 414. Meanwhile, in the angle path, the angle modulator 412 operates to modulate an RF carrier signal by the angle information in the analog angle modulation signal, thereby producing an angle-modulated RF carrier signal which is coupled to the RF input port RFin of the PA 414.
As shown in FIG. 5, the angle-modulated RF carrier signal at the RF input port RFin of the PA 414 has a constant envelope. This permits the PA 414 to be configured to operate in its nonlinear region of operation (i.e., as a “nonlinear” PA) without the risk of signal peak clipping, as was mentioned above, and the PA 414 can be operated without having to back-off the output power. Typically the PA 414 is implemented as a highly-efficient switch-mode PA (e.g., as a Class D, E or F switch-mode PA) switching between compressed and cut-off states. When configured in this manner, the amplitude-modulated power supply signal Vs(t) modulates the power supply port of the PA 414 and the envelope information is restored to the RF output RFout of the PA 414 as the PA 414 amplifies the angle-modulated RF carrier signal. By operating the PA 414 as a switch and dynamically controlling the power supplied to it, the polar transmitter 400 is able to achieve significantly higher energy efficiencies than the quadrature-modulator-based RF transmitter 200.
Although the polar transmitter 400 is able to handle non-constant envelope signals at higher energy efficiencies than the more conventional quadrature-modulator-based transmitter 200, the amplitude and angle component signals ρ and θ typically have much higher bandwidths compared to the rectangular-coordinate I and Q baseband signals from which they derive. This so-called “bandwidth expansion” phenomenon occurs during the rectangular-to-polar conversion process performed by the CORDIC converter 404. The high bandwidths are manifested as high-frequency events in the amplitude and angle component signals ρ and θ and are highly undesirable. Not only do the high-frequency events tend to degrade the modulation accuracy of the polar transmitter 400, they also cause the transmission spectrum to extend beyond its intended band-limited channel, resulting in adjacent channel interferers and an increase in receive band noise. These effects can be very difficult to deal with, especially when strict noise restriction standards must be adhered to.
The extent to which high-frequency events end up appearing in the amplitude and angle component signals ρ and θ is very much dependent on the modulation scheme that is employed. In particular, non-constant modulation schemes that produce signals having a high average-to-minimum magnitude ratio (AMR) generally have a very large angle component bandwidth. In fact, for modulation schemes that produce signals which pass through zero, as illustrated in the signal trajectory diagram in FIG. 6, phase changes by as much as 180 degrees can occur, resulting in an angle component signal θ having essentially infinite bandwidth. Signals of such high bandwidth cannot be accurately processed and transmitted by the polar transmitter 400, or by any type of transmitter for that matter, and the high-frequency content in such signals makes standards compliance extremely difficult, and in some cases impossible, to achieve.
Various techniques have been proposed to reduce high-frequency events in polar domain signals. One approach, known as “hole blowing,” involves identifying symbols (or samples of symbols) in the baseband signal s(t) during which the magnitude of the signal falls below a predetermined low-magnitude threshold, and then raising the magnitude of the baseband signal s(t) in the temporal vicinity of the identified symbols or samples so that the AMR of the signal is reduced. The term “hole blowing” is used since the effect of applying the technique is to produce a “hole” in the signal trajectory diagram of the baseband signal s(t). As illustrated in FIG. 7, the “hole” forces the trajectory of the modified baseband signal ŝ(t) to not pass too close to the origin, resulting in a desired reduction in the bandwidth of the signal.
The conventional hole blowing technique is described in U.S. Pat. No. 7,054,385 to Booth et al. As explained there, the baseband signal s(t) is modified by adding correction pulses to it, to form a modified baseband signal:
                    s        ^            ⁡              (        t        )              =                            ∑          n                ⁢                                  ⁢                              a            n                    ⁢                      p            ⁡                          (                              t                -                                  nT                  s                                            )                                          +                        ∑          m                ⁢                              b            m                    ⁢                      r            ⁡                          (                              t                -                                  t                  m                                            )                                            ,where r(t) is the correction pulse, m is the perturbation index, tm represents the times when the baseband signal s(t) is perturbed (i.e., the times when the correction pulse r(t) is inserted), and bm is a perturbation sequence representing the amplitude scaling and/or angle shifting applied to the correction pulse r(t).
As shown in FIG. 8, in generating the modified baseband signal ŝ(t) the baseband signal s(t) from the baseband modulator 102 is fed forward to an analyzer 802. The analyzer 802 then determines the perturbation times tm by detecting low-magnitude events in the baseband signal s(t) that fall below the fixed low-magnitude threshold. In response to detected low-magnitude events, the analyzer 802 generates the perturbation sequence bm. A pulse-shaping filter 804 generates the correction pulses r(t), scales the correction pulses by the perturbation sequence bm, and finally adds the scaled correction pulses to the original baseband signal s(t), to produce the desired AMR-reduced modified baseband signal ŝ(t).
The conventional hole blowing technique can be helpful in reducing the AMR of communications signals in polar transmitters configured to operate in accordance with some types of non-constant envelope modulation schemes. However, it does not always provide satisfactory results. Moreover, for multi-mode polar transmitters that support multiple non-constant envelope modulation schemes, which produce baseband signals having different AMRs, the conventional hole blowing technique is in most cases unacceptable. State-of-the-art multi-mode transmitters, such as those used in modern cellular telecommunications systems, are often designed to operate in both second generation (2G) and third generation (3G) mobile telecommunications systems. 2G mobile telecommunications systems employ Enhanced Data Rates for GSM (Global System for Mobile communications) Evolution (EDGE) to achieve higher data rates. GSM/EDGE uses an 8 phase shift keying (8PSK) non-constant envelope modulation scheme. 3G Wideband Code Division Multiple Access (W-CDMA) telecommunications systems also use non-constant envelope modulation schemes—Hybrid Phase Shift Keying (HPSK) and 16 quadrature amplitude modulation (16QAM), if High-Speed Packet Access (HSPA) protocols are used. The baseband signals generated from these different non-constant envelope modulation schemes all have different AMRs. Unfortunately, the conventional hole blowing approach described above, with its single, fixed low-magnitude threshold, is inadequate, and in many cases entirely incapable of, reducing the AMRs of the different baseband signals to levels necessary to guarantee compliance with the noise restriction specifications of all the various standards.
Long Term Evolution (LTE), the next generation mobile telecommunications technology, will also use a non-constant envelope modulation scheme known as Orthogonal Frequency Division Multiplexing (OFDM). When LTE is deployed, multi-mode transmitters will be designed to support OFDM as well as the non-constant envelope modulation schemes used in legacy W-CDMA systems. Problems associated with reducing the AMRs of signals in those and other next generation multi-mode transmitters will be similar to those encountered in present day GSM/EDGE and W-CDMA multi-mode transmitters.
Considering the drawbacks and limitations of conventional hole blowing approaches, it would be desirable to have methods and apparatus that are effective at reducing the AMR of communications signals in current and next generation communications transmitters.